LLC converters are widely used in the power stage of battery energy storage converters due to their excellent soft switching performance and low output impedance. Under low voltage and high current conditions, matrix transformers are often used on the secondary side of LLC to reduce current stress. In practical circuits, the volume of matrix transformers accounts for about 25% of the total main power, which seriously restricts the improvement of device power density. This paper proposes an integrated optimization design of LLC four-element-matrix planar transformer considering loss and parasitic parameters. Decoupling the influence of various structural parameters of transformers on parasitic parameters this method achieves efficient operation of transformers and controllable parasitic parameters. Voltage drop and oscillation are effectively suppressed.

Firstly, establish an accurate transformer loss model based on the proposed distributed magnetic core loss calculation method. Select the key parameters of the magnetic core that meet the efficiency and volume requirements: the radius of the magnetic core's central pillar r and the total width of the winding c. Afterward, select the winding layer structure. Establish a leakage inductance model based on transformer leakage magnetic field energy, determine the feasibility of leakage inductance integration through PCB thickness constraints, and design leakage inductance values.

The experiment shows that the secondary-side V_ds voltage oscillation is significantly reduced after integration optimization, reducing the parasitic capacitance value. By matching the resonant inductance to ensure that LLC operates in critical continuous mode at 300 kHz, the magnitude of the transformer's primary leakage inductance before and after integration can be calculated. The resonant inductance before integration is 1.8 μH, indicating the original edge leakage is 0.7 μH. After integrated optimization, the resonant inductance is 2.3 μH, indicating the original edge leakage is 0.2 μH. When operating in reverse, with the same input voltage of 3.2 V, the LLC output voltage rises from 34 V to 35 V. This means that the secondary edge leakage has decreased after integration. After resonance point matching, the secondary leakage inductance can be reduced from 33 nH to 12 nH. The secondary side V_ds oscillation caused by parasitic capacitance is close and small, which verifies the control effect of parasitic parameters.

This method can achieve the following effects. (1) The proposed distributed magnetic core loss calculation method for planar transformers based on P-B curves eliminates the influence of uneven magnetic density distribution on the accuracy of the magnetic core loss model, achieving accurate modeling of integrated transformer losses. (2) This method provides a judgment method for the feasibility of leakage inductance integration. (3) The prototype parasitic parameters can be controlled by accurately modeling the leakage inductance and parasitic inductance. This method provides theoretical support for designing and optimizing energy storage converters in LLC low-voltage and high-current scenarios.

With the remarkable growth of renewables, distributed power generation systems (DPGSs) are starting to take over the dominant role of synchronous machines. As an essential interface between renewables and power grids, the grid-connected inverter plays an important role in the safe and stable operation of DPGSs. Among two types of grid-connected inverters, i.e., grid-following (GFL) and grid-forming (GFM) ones, attention has gradually turned to the GFM inverter in recent decades, owing to its synchronous-machine-like characteristics and capability of operating in weak grid or even forming a stand-alone grid. However, similar to the GFL inverter, the GFM inverter may exhibit non-passive characteristics in the mid/high-frequency bands, leading to mid/high-frequency resonance risk.

The existing research mainly focuses on sub-synchronous oscillation, but the mid/high-frequency resonance issue still needs to be explored. In order to mitigate the mid/high-frequency resonance and harvest the desired performance, this paper provides the optimal design procedure for controller parameters from the perspective of internal stability and the impedance reshaping method via the grid current feedforward from the perspective of external stability.

Firstly, a mathematical model of the voltage-current double-loop controlled GFM inverter is established. The control block diagram of the inverter’s control system is depicted, and its equivalent transformation is performed. Accordingly, the impedance model of the GFM inverter is obtained as a controlled source in series with the output impedance.

After that, the stability of the GFM inverter is divided into internal stability and external stability, which characterize the stability of the equivalent voltage source and the interaction stability between the equivalent impedance and the grid, respectively. From these two stability dimensions, the stability mechanism and resonance risk of the GFM inverter are analyzed based on the Nyquist stability criterion and the passivity theory, and the main factors affecting the system stability are revealed.

According to the internal stability constraints, stability margin requirements, and steady-state error, an optimal design procedure for the control parameters is provided, which avoids repeated trials and ensures internal stability and low steady-state error. Additionally, based on the external stability constraints, the impedance shaping scheme with the grid current feedforward is proposed, and the corresponding feedforward function is derived. The proposed scheme is simple to implement and can effectively enhance the inverter's robustness against grid impedance variations.

Finally, experiments are carried out on a 10 kW GFM inverter prototype. The results confirm that under different grid conditions, the inverter can with the designed parameters and the proposed scheme continuously operate stably, and the power quality is high, which verifies the theoretical analyses and the proposed scheme.

The high peak-to-average ratio of low-frequency pulse power loads seriously affects the safe and stable operation of airborne power supply systems. The conventional approach requires stacking numerous energy storage capacitors due to the DC bus voltage ripple limitation, which substantially increases the system’s volume. Although current active pulse power suppression method can reduce the required capacitance by increasing voltage fluctuations, the considerable power ratings and additional power processing stages of active suppression circuits impact system efficiency significantly.

This paper presents a low-frequency pulse power active suppression method based on voltage compensation. The active suppression circuit is inserted between the DC bus and the energy storage capacitor C_d. By compensating for the voltage difference between C_d and DC bus with the output voltage v_s of the active suppression circuit, the voltage range of C_d is not constrained by the DC bus, allowing for a reduction in C_d. Since the active suppression circuit only compensates for capacitor voltage fluctuations, its power rating and losses are much smaller than the average power of pulse loads, which greatly reduces the volume, weight, and losses. The active suppression circuit takes power from the DC bus. Considering that its input and output terminals are non-common ground and have a wide output voltage range, the LLC-DC transformer (DCX) cascaded Buck converter is chosen for the active suppression circuit. The LLC-DCX functions operate in an open loop as a high- frequency DC transformer, and a dual-loop control strategy is implemented for the Buck converter. The outer voltage loop adjusts the voltage fluctuation range of C_d, while the inner current loop suppresses the input current ripple.

The design guidelines for key parameters are also presented, with size and efficiency as the main considerations. The size of the power supply is influenced by C_d, while the power rating of the active suppression circuit affects system efficiency. Therefore, a detailed study of both aspects is conducted. The results reveal that once the load is determined, C_d decreases as the voltage fluctuation Δv_d increases, and the decreasing rate gradually slows. Additionally, the power rating of the active suppression circuit increases linearly with the average voltage V_dav and Δv_d. The lower limit of V_dav is also affected by Δv_d to ensure that the output voltage of the Buck converter remains positive. Therefore, a balance must be achieved between V_dav and Δv_d to optimize capacitance and power rating.

An experimental prototype is constructed. The experimental results are consistent with the theoretical analysis, and the active suppression circuit effectively regulates the voltage fluctuation range of C_d and suppresses the input current. Efficiency tests reveal that the active suppression scheme maintains an efficiency above 98.1% throughout the entire range, with a peak efficiency reaching 99.1%. This scheme is compared with existing active suppression schemes, showing clear advantages. In addition, results from various literature are normalized and compared.

Traditional control parameter design methods of grid-connected converters (GCCs) are usually carried out under rated operation conditions. The stability margin is generally characterized by the magnitude margin (GM) and phase margin (PM). The bandwidth of the phase-locked loop (PLL) needs to be sacrificed for enough system stability margin, which ensures that the system can operate safely under non-rated working conditions. Therefore, traditional parameter design methods make it difficult to achieve a compromise between the stability and rapidity of GCC. Furthermore, the relationship between the stability margin and the output limit of the system is difficult to obtain, which relies on simulations or experiments. To address this issue, this paper proposes a control parameter design method based on the system stable operation domain, which com- prehensively considers the variable working conditions and different stability requirements.

Firstly, the complex vector open-loop transfer function G_s(s) is derived, which can analyze the stability of the system in a wide operating range or under different parameters. Secondly, varying the values of the PLL and current loop control parameters, the feasible domain of control parameters can be obtained by numerical analysis. PLL and current loop are coupled to each other due to the grid impedance. Therefore, the phase-locked loop parameter has a stable parameter boundary. Thirdly, based on the upper limit of the obtained PLL parameter, the three-dimensional diagram between the PLL parameter and the output current of the dq-axis is plotted, which shows the stable operation boundaries of GCC under different PLL parameters. Finally, taking the operation margin as the stability margin, the PLL parameter can be flexibly designed to realize the balance between the stability margin and the dynamic performance. Theoretical analysis results show that the control parameter design method based on the stable domain can ensure the safe and reliable operation of the system. Compared with traditional parameter design methods, determining the value of control parameter based on operation margin can improve the dynamic performance of PLL. At the same time, the corresponding relationship between different stability margin and the output current limit of the system can be obtained. Finally, the simulation and experiment results verify the correctness of the theoretical analysis and the effectiveness of the proposed design method.

The following conclusions can be drawn from the theoretical analyses: (1) The derived open-loop complex transfer function can be used to plot the control parameter feasible domain and the stable operation domain of GCC. The PLL control parameter satisfying the operation margin can be obtained quickly without repeated trial and error in the parameter design. (2) The analysis results show that the grid strength is positively correlated with the parameter feasible domain and the PLL cutoff frequency is negatively correlated with the stable operation domain. The cutoff frequency value of PLL is limited by the current loop parameter. The smaller the operating current in the parameter design, the wider the range of PLL cutoff frequency, and the better the dynamic performance of the PLL. (3) The control parameter design method based on the stable operation domain can directly quantify the influence of stability margin on the limit of the output current, which can be used for obtaining the stable boundaries under different stability margin. At the same time, the control parameters of PLL can be flexibly designed according to the actual needs of rapidity.

The magnetic properties and loss characteristics of oriented silicon steel sheets exhibit significant deviation under stress. The traditional loss separation model generally overlooks the impact of mechanical stress on the loss characteristics, resulting in calculation errors. In recent years, most studies on the loss characteristics of oriented silicon steel sheets under mechanical stress have focused on qualitative analysis, with only a few studies making quantitative improvements to the loss separation model. This paper develops an improved loss separation model based on the traditional loss separation model by introducing stress terms into the hysteresis loss and excess loss.

Firstly, measurements from a single sheet tester with unidirectional stressing are utilized to analyze the stress dependency of the loss characteristics of the oriented silicon steel sheets. The experimental results demonstrate a significant enhancement in loss under compressive stress while exhibiting a slight decreasing trend under tensile stress. The magnetization mechanism in ferromagnetism explains the variation of the loss characteristics under mechanical stress. Secondly, the hysteresis loss and excess loss under stress are calculated based on the Bertotti traditional loss separation model. Since the stress component is not introduced into the hysteresis loss in the traditional loss separation model, the effect of stress on the hysteresis loss is only reflected by the hysteresis loss coefficient, leading to a significant error in the calculation of the hysteresis loss under stress. Although the excess loss parameter ${{V}_{0}}$, currently expressed by a constant coefficient, embodies the effect of stress, it fails to capture the effect of the applied mechanical stress on the losses of each magnetic induction intensity. Consequently, computational inaccuracies arise when employing the Bertotti traditional loss separation model.

Based on the correlation between parameters ${{V}_{0}}$ in excess loss, hysteresis loss, and stress, the traditional separation formula for losses is improved by introducing stress components into the excess loss parameters ${{V}_{0}}$ and hysteresis loss. An improved loss separation model is established and verified by varying the frequency of excitation and the type of oriented silicon steel sheet. The results indicate that the improved loss separation model can accurately separate and calculate the losses of oriented silicon steel sheets under different stresses while maintaining a remarkable precision level.

Experimental measurement and calculation analysis are performed, and the conclusions are as follows. (1) The excess loss parameter ${{V}_{0}}$ is correlated with stress, and incorporating the stress component into the excess loss parameter can effectively mitigate the calculation error caused by stress in the traditional loss separation model. (2) An improved loss separation model is proposed based on the traditional mode by incorporating the excess loss and hysteresis loss into stress-related functions. (3) The improved loss separation model is confirmed through testing with different frequency excitations and oriented silicon steel sheets, demonstrating its ability to accurately separate losses under different stresses.

The dynamic transformer rating (DTR) and thermal life loss of oil-immersed transformers under on-site variable load operation conditions are closely related to the transient temperature rise of the equipment. However, as an implicit solution method, the traditional finite element analysis and finite volume method need to be iteratively solved in each sub-step of transient thermal analysis, which has many computational resources and is time consuming. It is challenging to meet the requirements of fast calculation. The rapid and accurate solution of the temperature field (especially the hot spot temperature rise) of the oil-immersed transformer in field operation is the premise to realize the digital operation and maintenance of the transformer and the DTR evaluation. Therefore, this paper proposes a lattice Boltzmann (LBM) physical in the loop simulation model for coupled electrical networks. The real time evaluation of DTR under electrical network constraints is realized through the rapid solution of the transformer temperature field.

Firstly, the D2Q9 model is used to solve the fluid flow and thermal lattice Boltzmann equations (LBEs) to capture the transient oil flow and temperature rise process inside the transformer. In the Simulink environment, the equivalent current source model is used to construct the electrical network constraints of multi-level load scenarios, and the established transformer LBM model is used as a component for numerical encapsulation to complete the construction of the physical-in-the-loop simulation model. Secondly, to verify the effectiveness of the proposed method, the finite volume method (FVM) is used to simulate the same oil-immersed transformer model. The grid independence test determines the optimal number of grids. The number of grids is 1 250×420 for LBM modeling and simulation, and the number of units is 43 654 for FVM meshing. Compared with the constructed LBM-Simulink model with the FVM model, LBM still has the advantages of speed and memory occupation when the number of lattices is higher than the number of FVM units. If commercial software is used, this advantage will be further expanded. Thirdly, the steady state solution results of the hot spot temperature rise of the established LBM model are compared with the FVM solution, and the error is 2.60%. According to the load curve given in the transformer guidelines, it is used as input to solve the transient temperature rise of the established LBM and FVM simulation models. Finally, the results show that the LBM and FVM calculations are better than the transformer guide calculation. The maximum error between the hot spot temperature calculated by LBM and FVM is 6.44%. Moreover, the hot spot temperature rise trend of LBM is consistent with the transformer load guidelines, which verifies the effectiveness of the proposed method.

Based on the constructed LBM model, the load capacity of oil-immersed transformers under constant 25℃ and typical ambient temperature changes in summer and winter are evaluated at 6~18 hours during the day. The results show that under the premise that the relative insulation life loss of oil-immersed transformers is less than 1. The maximum load capacity coefficients are 1.20, 1.10, and 1.60 under the constant ambient temperature of 25℃, typical temperature changes in summer, and typical temperature changes in winter. The simulation model based on the proposed LBM provides an effective method for real-time monitoring of temperature rise, load capacity evaluation, and dynamic capacity increase of oil-immersed transformers.

The magnetic lift control rod drive mechanism (CRDM) is a critical electromagnetic actuator for regulating nuclear reaction rates. Its dynamic process is complex to predict due to the cross-coupling among current response, magnetic circuit saturation, and motion state. The latest equivalent magnetic network (EMN) model exhibits inaccuracies and relies on flux distribution during modeling, lacking generality. In multi-field coupling analysis, researchers often employ multi-software collaborative or semi-simulation methods, which incur significant time and hardware costs. This paper proposes a dynamic equivalent magnetic network (DEMN) model and a multi-physics field coupling calculation method considering transient current changes and saturation.

Firstly, the structure of the magnetic lift CRDM is introduced, and its lift solenoid valve is selected to analyze the electromagnetism-mechanics coupling during dynamic processes. Then, the mechanism is partitioned using orthogonal grid lines, and mesh units of multiple media are consolidated into a single medium to unify the reluctance calculation formula. During dynamic changes, only the grid size or position in the motion region is altered, thereby eliminating redundant modeling and reducing computational errors caused by mesh discrepancies. A connection relationship and calculation method for branch reluctance are established to address the misalignment between fixed and moving mesh units, facilitating continuous armature movement. A multi-physics field coupling calculation method is proposed by combining circuit models and kinematic formulas, which consider transient current changes and saturation. Finally, Compared with 3D finite element analysis (FEA) and experiments, the DEMN model and the proposed multi-field coupling calculation method are verified.

3D FEA results show that the magnetic density distribution, inductance, and electromagnetic force are highly consistent with the DEMN results, where the maximum error of inductance is 5.7%, and the maximum error of electromagnetic force is 3.4%. The experiments show that linear and saturated inductance variations are similar. The calculation accuracy of the release and suction currents under various loads exceeds 91%. Additionally, the dynamic results closely align with experimental results, with motion time calculation errors at 28 A and 40 A currents of 0.99% and 2.99%, respectively.

The conclusion is as follows. (1) By using orthogonal grid lines, the unified calculation of reluctance is achieved, accelerating the modeling speed. (2) By changing the grid size or position of local areas, the dynamic changes of the EMN model are achieved, avoiding the problem of repeated modeling in the multi-field coupling calculation process and reducing the calculation errors caused by differences in mesh partitioning. (3) Compared with FEA, the proposed DEMN model requires less computational resources and shorter computation time while ensuring accuracy. Compared with the experimental results, the effectiveness and accuracy of the DEMN model and multi-field coupling calculation method are verified. It can be extended to EMN modeling, performance analysis, and rapid optimization design of the entire CRDM.

In non-destructive testing (NDT), there is a growing demand for simulation tools that can predict magnetic characteristics, enhance understanding, and avoid harsh and uncertain experimental expectations. Due to the high sensitivity and non-destructive nature, the measurement and simulation of magnetic barkhausen noise (MBN) have become important in NDT.

This paper measured the MBN signals of soft magnetic materials under different stress conditions at a magnetic frequency of 10 Hz. The experimental results revealed the significant impact of tensile and compressive stresses on the MBN signals. Specifically, as tensile stress increases, the spacing between magnetic domain walls decreases, reducing energy loss in the movement of domain walls. The migration rate of the domain walls is accordingly increased, which in turn causes the MBN signals to rise. At the same time, due to the presence of additional domains in oriented silicon steel, the MBN signals exhibit a double-peak structure. As tensile stress increases, these additional domains are suppressed. Hence, peak-to-peak values one and two increase, and the increase in peak value two is significant. When a magnetic field and compressive stress are applied along the rolling direction, the compressive stress increases the energy of the magnetic domain walls, reducing their migration rate and weakening the MBN signals. Ithelpsto better understand the changes in the magnetic properties of soft magnetic materials under different stress conditions.

Existing MBN models can not accurately simulate the MBN signals of different soft magnetic materials under stress. This paper proposes a mathematical model based on the improved S-J-A hysteresis model. This model simulates MBN signals by considering the irreversible motion of magnetic domain walls in soft magnetic materials, thereby increasing the accuracy of the simulation. Specifically, the improved S-J-A hysteresis modelsimulates the irreversible hysteresis loops of soft magnetic materials, considering the relationship between magnetic anisotropy, model parameters, and stress. Then, these irreversible hysteresis loops are linked with the MBN envelope line to establish a mathematical model for the MBN envelope curve. Next, the MBN signals are simulated by modulating white noise in the 1~50 kHz range with this envelope curve. Finally, three different soft magnetic materials are selected: oriented electrical steel sheet (30QG120), non-oriented silicon steel (35WW230), and amorphous alloy (1K101). The proposed MBN model simulates MBN signals under different mechanical stress conditions.

The proposed MBN model accurately simulates the MBN signals of the oriented electrical steel sheet (30QG120), non-oriented silicon steel (35WW230), and amorphous alloy (1K101) under stress. A comparison of MBN signals between oriented silicon steel, non-oriented silicon steel, and amorphous alloy is conducted, revealing that the double-peak structure exhibited by oriented silicon steel under tensile stress is related to its anisotropy. Microscopic analysis gains a deep understanding of the stress effects on the magnetism of soft magnetic materials and the generation mechanism of MBN. The proposed MBN model provides a reliable tool in material characterization and non-destructive testing (NDT) applications, laying the foundation for further engineering applications.

Epoxy resin (EP) possesses advantages such as low cost, high mechanical strength, robust chemical resistance, and excellent electrical insulation properties, making it extensively utilized in various epoxy cast electrical equipment like dry transformers and reactors. Nonetheless, the three-dimensional cross-linked network of resins exhibits non-melting characteristics, posing challenges in the degradation and recycling of retired epoxy electrical equipment. Nowadays, researchers have achieved epoxy resin recycling by incorporating dynamic covalent bonds into the epoxy resin crosslinking network to develop degradable Vitrimer epoxy resin materials. However, when applied to complex environments like high temperature, humidity, and intense electric fields, the internal crosslinking network of epoxy Vitrimers material may deteriorate, impacting its operational longevity. Therefore, besides ensuring favorable electrical, mechanical, and thermal properties, the enduring reliable performance of Vitrimer resin cannot be disregarded. This paper prepared dual-dynamic bonds Vitrimer resin with varying disulfide bond contents. The micromorphology, electrical characteristics, mechanical properties, dynamic thermodynamic properties, and degradation properties of Vitrimer resin at diverse aging stages were regularly investigated and the life evaluation model is constructed at last.

Firstly, dual dynamic crosslinked Vitrimer resin basded on ester bonds and varied disulfide bonds were prepared with 3,3’-dithiodipropionic acid (DTDPA) and hexahydro-4-methylphthalic anhydride (MHHPA) as the curing agent, with Triethanolamine acting as the catalyst. Subsequently, the accelerated thermo-oxygen aging tests were carried out, during which the microscopic morphology, electrical properties, bending characteristics, dynamic thermodynamic attributes, and degradation properties of vitrimer resin were periodically evaluated. Experiment results revealed alterations in the resin's microstructure under hot oxygen aging, leading to random internal cross-linked network fractures that generate abundant free radicals, ultimately causing resin failure. The resin's bending strength diminishes, rigidity increases, toughness notably decreases, and the bending fracture transitions to a brittle fracture pattern. As aging progresses, a denser cross-linked network forms on the resin's surface, elevating T_g. The integration of disulfide bonds makes the resin system more susceptible to oxidation and molecular chain breakage, resulting in reduced breakdown voltage, heightened dielectric loss factor, and increased insulation deterioration. Throughout the aging process, the degradation rate of Vitrimer resin in glycol solution decreases due to surface ester bond reduction and oxide layer formation, while the destruction of the disulfide crosslinking network prevents resin degradation in dithiothreitol solution. Lastly, a life evaluation model for the dual dynamic crosslinked Vitrimer resin was formulated based on the results of bending strength and TGA tests.

The dual dynamic crosslinked Vitrimer resin has excellent comprehensive properties and can realize the recycling of decommissioned epoxy electrical equipment. In this paper, the effect of thermal oxygen aging on the properties of dual dynamic crosslinked degradable resin was studied, which laid the experimental and theoretical foundation for the long-term service of vitrification epoxy resin in electrical equipment.

For bundled conductors, the shadowing effect of upwind sub-conductor will affect the airflow and droplets distributions of downwind one, resulting in the difference in icing characteristics. Traditional icing calculation process generally ignored these differences and, hence, only giving an identical icing mass result of each sub-conductor. This affects the study of the aerodynamic characteristics and deicing methods of icing bundled conductor. Although some scholars have pointed out that the shadowing effect between sub-conductors will influence the icing process, there is no quantitative study. Therefore, this paper further explores the shadowing effect and relevant influencing factors of bundled conductors through numerical simulation and test research. Furthermore, based on the analysis of the shadowing effect and the superposition principle, a rapid calculation method of ice mass accreted on the bundled conductor is proposed.

Firstly, the distributions of airflow and droplets around bundled conductor are solved by Eulerian-Eulerian two-phase flow model. Secondly, combined with the mass and thermodynamic balance equations, the icing mass and shape accreted on bundled conductor under various icing environments are obtained. Then a new parameter called shadowing coefficient is defined to investigate the shadowing effect and influencing factors as well. The results show that: Shadowing effect is weakened with increasing absolute value of shadowing angle and bundled-spacing, but intensified with the increase of median volume diameter (M_VD) of droplets; Meanwhile, the shadowing effect experiences a growth and then drops down along with the increase of wind speed, and reach to the max at 15 m/s range 5~20 m/s.

Based on the superposition principle and shadowing effect analysis, a rapid calculating method for ice mass on bundled conductor is proposed. Where iced bundled conductor is regarded as a linear combination of non-shadowed sub-conductor (single conductor) icing intensity and shadowing coefficient, so the icing intensity of various types of bundled conductor can be obtained only requiring the icing intensity of single conductor and the shadowing coefficient in the corresponding environment. Then the rapid icing calculation formulars of 3,4,6,8-bundled conductor under various shadowing angle is given by geometry analysis, respectively, which simplifies the calculation of the icing mass on bundled conductor.

Finaly, a 4-bundled conductor nature icing test was carried out at the Xuefeng Mountain Energy Equipment Safety National Observation and Research Station to validate the accuracy of the numerical simulation and rapid calculation method. Results show that under the environment parameters of ambit temperature T_f = -2℃, M_VD = 25.4 μm, liquid water content L_wc = 0.61 g/m³, wind speed V = 10 m/s and shadowing angle θ = 2°, the difference in icing intensity between rapid calculation and test results was within -4.01% to -19.77%, the icing thickness differences of sub-conductors were between 1.66% to -6.36% and the differences in shadowing coefficient were between 4.05% to 5.33%, which well verifies the accuracy of the rapid calculation method proposed in this paper.